Research Article  Open Access
Hongzhu Li, Ling Zhu, Le Wang, "A DoubleBoost Converter Based on Coupled Inductance and Magnetic Integration", Active and Passive Electronic Components, vol. 2021, Article ID 8014620, 15 pages, 2021. https://doi.org/10.1155/2021/8014620
A DoubleBoost Converter Based on Coupled Inductance and Magnetic Integration
Abstract
Highvoltage gain converter has a highfrequency use in some industrial fields, for instance, the fuel cell system, the photovoltaic system, electric vehicles, and the highintensity discharge lamp. In order to solve the problem of the lowvoltage gain of traditional boost converter, the doubleboost converter with coupled inductance and doubled voltage is proposed, which connects the traditional boost converter in parallel. The voltage gain of the converter is further improved by introducing the voltagedoubled unit of the coupled inductance. Moreover, the clamp capacitor can absorb the leakage inductance in the circuit and reduce the voltage stress of the switch. In addition, two coupled inductors are magnetically collected; then, the loss of the core is analyzed under the same gain. The detailed analysis of the proposed converter and a comparison considering other topologies previously published in the literature are also presented in this article. In order to verify the proposed converter performance, a prototype has been built for a power of 200 W, input and output voltages of 12 and 84 V, respectively, and a switching frequency of 50 kHz. Experimental results validate the effectiveness of the theoretical analysis proving the satisfactory converter performance, whose peak efficiency is 95.5%.
1. Introduction
The output voltage of the photovoltaic circuit board is low. A firstlevel DC converter should be used to connect the grids, which can set the output voltage of the photovoltaic circuit board to reach the voltage requirement of the gridconnected inverter. Meanwhile, due to the influence of light intensity, the output voltage range of the photovoltaic circuit is quite large, so highgain DC converters have always been one of the research hotspots at home and abroad [1–4].
In addition, in order to achieve a highvoltage gain in a conventional boost converter, a high duty cycle should be used, which may result in high power loss. Most converters are required to operate at a higher duty cycle and a higher number of component counts, making it easier to saturate the topology with inductive cores. This puts limitations on the converter. Literature [5] proposed an optimized integration of active switching inductor configuration, trapezoidal voltage doubling unit combined with regenerative boost configuration, suitable for DCDC converter with a wide range of voltage gain in low duty cycle range, because the input clamped peak current can achieve higher voltage gain at a low duty cycle. In literature [6], switching inductance, switching capacitance, and coupled inductance techniques are implemented using threewinding coupled inductance to achieve highvoltage gain with low turns ratio. With the use of a lowvoltage stress power switch, through the recovery of leakage energy, the voltage peak on the power switch can be well suppressed so as to improve the efficiency of the system. The proposed converter in literature [7] is composed of a double switching structure, a threewinding coupling inductor, and two voltage multipliers. The double switching structure is beneficial to reduce the voltage stress and current stress of the switch. Clamping capacitors recover energy from leakage inductors, and the problem of inductor leakage and diode reverse recovery is alleviated, improving efficiency. A novel noncoupled high stepup DCDC converter is presented in literature [8, 9], which combines passive switched capacitance (PSC) network (also known as 2cell step (SU2C) with ASL network. The ASLSU2C converter is formed. The current output feature of ASLSU2CCO (which is called in the article ASLSU2CCO) and the voltage output (VO) feature of ASLSU2CVO are studied, respectively. ASLSU2CCO uses fewer switching devices to effectively reduce losses and improve efficiency. There are only two power switches on the current path, and the voltage stress on the active switch is lower. ASLSU2CVO has the characteristics of ASLSU2CCO and can provide higher static gain, while its switching is subjected to lower voltage stress, which can be better applied in solar and smallscale wind energy systems, storage energy systems, and other applications requiring nonisolated highorder converters. In literature [10], a novel DC/DC converter combining active switch and coupled inductor structure is proposed. The ASL structure reduces the current stress on the switches during the charge interval, the active clamp technology provides zerovoltage transition (ZVT) for all switches, and the leakage inductance can alleviate the reverse recovery problem of the output diode. Literature [11] proposed that the first coupled inductorbased converter employs twodiode capacitor voltage multipliers (VMs) inserted in an innovative way to give it many advantages. The first involves VMs located near the switch and an output circuit, and two VMs work together to provide a very large DC gain. The second method combines forming a new hybrid VM cell with a single VM. The recovery problem of the output diode is eliminated. No spike occurs at its turnoff. Its advantage is eliminating the output diode recovery problem and thus does not result in spikes at its turnoff. In literature [12], a voltage multiplier unit and a switching capacitor unit are combined to achieve low voltage and low current stress efficiency between components. In literature [13–17], a threephase asymmetric inductor was used to replace a threephase discrete inductor and an integrated magnetic structure technology was adopted. This reduces the total volume of the core, thus increasing the efficiency of the system. Literature [18] connected the basic boost converter and the voltagedoubled unit in parallel. All stages are activated by the boost switch. Each converter contains a switched capacitor circuit, which has a simple structure and can recover leakage inductance energy. In literature [19], the traditional boost converter and SEPIC converter are used as an alternative structure of the series output module. The boost converter and the SEPIC converter use the same boost inductor and the same switch, which can reduce the input current ripple while maintaining the characteristics of the boost converter. The circuit topology proposed in [20] uses switched capacitor voltage doubler and an integrated LC2D output network to increase the voltage gain of the converter and reduce the voltage stress on the power switch. It has a continuous input current, lowvoltage stress on its semiconductor device (less than half of the output voltage), and a constant potential difference between its input and output grounded terminals. The converter is under a wide range of voltage gain and can adapt to the soft output characteristics of the fuel cell. This research proposes a dual boost converter with coupled inductance voltage doubling unit clamped to the ground. In order to effectively improve the voltage gain of the converter, two traditional boost converters are combined in parallel, the clamp capacitor reduces the voltage stress of the switch tube, and the latter stage of the converter adds a coupled inductance voltage double unit composed of a coupled inductor, a diode, and a capacitor, which improves the gain and reduces the voltage stress of the switch tube. Analysis of the topological structure and working principle of the converter is quite detailed and exhaustive. The theoretical analysis shows that the gain of the converter is related to the turns ratio and the duty cycle of the coupled inductor. The coupled inductor is integrated together. Under the same voltage gain, a different duty cycle is obtained by changing the original and secondary side turns ratio n of the coupled inductor. The loss of the magnetic core under different duty cycles is analyzed to obtain the highest efficiency of the converter output. A practical 200 W prototype is built to verify the correctness of the theoretical analysis.
2. Topological Structure and Working Principle of the Converter
2.1. Topological Structure
The topological structure of the converter is shown in Figure 1. Two traditional boost converter topologies are connected in parallel at the input end, and the same structure is combined. The output ends are superimposed in a series to form the front stage of the output converter. The introduction of a coupled inductance voltagedoubled unit in the backstage of the converter further improves the outputinput voltage ratio, forming a highvoltage gain boost converter, as shown in Figure 2. On this basis, a clamping capacitor is introduced to absorb the leakage inductance in the coupled inductor and suppress the reverse problem of the diode.
For the convenience of the analysis, the following assumptions are proposed:(1)All switching tubes and diodes are ideal devices, both switching tubes and diodes are ideal devices, and the conduction voltage drop is zero(2)The coupled inductor is an ideal device, ignoring its parasitic parameters(3)The capacitor in the converter is an ideal device, and its capacitance value is large enough to ignore capacitor voltage fluctuations
2.2. Working Modes of the Converter
The equivalent circuit of the converter is shown in Figure 3. In a switching cycle, the CCM (continuous conduction mode) of the converter has five working modes, and the working waveforms of each mode are presented in Figure 4. Different working modes of each converter have different equivalent circuit diagrams, as shown in Figure 5:(1)Mode I [t_{0}−t_{1}]: from t_{0}−t_{1}, the switches S_{1} and S_{2} are closed; the diodes D_{1}, D_{2}, and D_{4} are off, and D_{3} and D_{5} are on. The equivalent circuit is shown in Figure 5(a). The excitation current starts to rise linearly under the action of the power supply V_{in}, and the secondary side currents i_{L1}, i_{L2} present a linearly decreased trend. The power supply, along with the primary and secondary sides of the coupled inductor and the capacitors C_{1}, C_{2}, C_{3}, C_{4}, provides power to C_{5} and the load together. When the secondary side currents i_{L1} and i_{L2} drop to 0, the diodes D_{3} and D_{5} are cut off, and mode I ends.(2)Mode II [t_{1}−t_{2}]: from t_{1}−t_{2}, the switches S_{1} and S_{2} are closed; the diodes D_{1}, D_{2}, D_{3}, and D_{5} are off, and D_{4} is on. The equivalent circuit is presented in Figure 5(b). The power supply continues to store energy, and that energy goes to the primary side of the coupled inductor. The leakage current and the excitation current rise linearly. The power supply, along with the secondary side of the coupled inductor and also the capacitors C_{1}, C_{2}, C_{3}, C_{4}, forms a closed loop, and the capacitor C_{5} supplies energy to the load alone. Leakage inductance current i_{Lk} shows a linearly rising trend, and when reaching the maximum state, mode II ends.(3)Mode III [t_{2}−t_{3}]: from t_{2}−t_{3}, the switches S_{1} and S_{2} are off, and the diodes D_{1}, D_{2}, and D_{4} are on. The equivalent circuit is presented in Figure 5(c). The leakage inductance current is reversed, which will cause a linear decrease of the current on the primary side of the coupled inductor. Capacitors C_{1} and C_{2} can absorb the leakage inductance in the circuit through diodes, which will result in reducing the voltage stress at both ends of the switch tube; the power supply, the secondary side of the coupled inductor, and also the capacitors C_{1}, C_{2}, C_{3}, C_{4} together establish a closed loop. As a result, the secondary side of the current of the coupled inductor will gradually decrease. The capacitor C_{5} supplies energy to the load alone. When the secondary side current drops to 0, diode D_{4} is off while D_{5} is on, and mode III ends.(4)Mode IV [t_{3}−t_{4}]: from t_{3}−t_{4}, the switches S_{1} and S_{2} are off; the diodes D_{1}, D_{2}, D_{3}, and D_{5} are on, while D_{4} is off. Figure 5(d) shows the equivalent circuit. The leakage inductance current L_{k} presents a linearly decreasing trend, as it is consistently absorbed by the capacitors, both C_{1} and C_{2.} The power supply, the primary and secondary sides of the coupled inductor, the diodes D_{1}, D_{5}, and the capacitor C_{3} together charge C_{5} and power the load; when the current of diodes D_{1}, D_{2} drops to 0, the diodes D_{1}, D_{2} are off, and modal IV ends.(5)Mode V [t_{4}−t_{5}]: from t_{4}−t_{5}, the switches S_{1} and S_{2} are off; the diodes D_{1}, D_{2}, and D_{4} are off, and D_{3} and D_{5} are on. The equivalent circuit is shown in Figure 5(e). At this time, the capacitors C_{1} and C_{2} are equivalent to the power supply. The power supply and capacitors C_{1}, C_{2}, and C_{3} supply energy to the load through the diode D_{5}. When the switches S_{1} and S_{2} are on, mode V ends.
(a)
(b)
(c)
(d)
(e)
3. Performance Analysis
3.1. Voltage Gain Analysis
In order to facilitate calculation and analysis, set , , The coupling coefficients of the two coupled inductors are both , and the coupling coefficient can be expressed as follows:
Assuming that the turns ratios of the coupled inductors are the same, that is, n, the turns ratio n can be expressed as follows:
The short transition modes I, III, and V are ignored in order to better analyze the steady state. Only modes II and IV are considered here.
In mode II, the loop voltage equation is
In mode IV, the loop voltage equation is
On the grounds of the voltsecond balance principle of the coupled inductor L_{m}, the following expression of an integral equation is obtained:
According to equations (1) and (3)–(5), the voltage expression of capacitors C_{1} and C_{2} is obtained:
According to equations (1) and (3), the voltage expression of capacitors C_{3} and C_{4} is obtained:
According to equations (4), (6), and (7), the voltage gain expression of the converter working in CCM mode is obtained:
According to equation (8), there is a strong relationship between the actual gain of the converter, the coupled inductance turns ratio n, and the coupled coefficient k. The relationship curve is presented in Figure 6, which shows that the greater the turns ratio n of the coupled inductor, the more significant the effect of the converter voltage gain. The increase of the leakage inductance L_{k} will result in a reduction to the coupled coefficient k. In that case, the voltage gain of the converter presents a decreasing trend, which will bring a certain loss of the duty cycle. So in practical applications, the primary and secondary sides of the coupled inductor should be as tightly coupled as possible. Therefore, the higher the coupling degree, the better when the turns ratio is the same.
When k = 1, the voltage gain of the converter is as follows:
The voltage stress of the switch tubes S_{1} and S_{2} is
The voltage stress of diodes D_{1} and D_{2} is
The voltage stress of diodes D_{3} and D_{4} is
The voltage stress of diode D_{5} is
According to formulas (10)–(12), the threedimensional curved surface diagram of the ratio of voltage stress at both ends of switches S_{1}, S_{2} and diodes D_{1}, D_{2}, D_{3}, D_{4} to output voltage V_{vps}/V_{o}, the coupled inductance turns ratio n, and duty cycle D is shown in Figure 7. It can be seen from Figure 7 that the voltage stress of all power devices is less than the output voltage under different duty cycle D and different turns ratio n. With the increase of turns ratio n, the voltage stress of switch tubes S_{1} and S_{2} decreases gradually, the voltage stress of diodes D_{1} and D_{2} decreases slowly, and the voltage stress of diodes D_{3}, D_{4}, and D_{5} increases slowly.
According to the analysis in Section 1 above, the voltage stress of the capacitors C_{1}, C_{2}, C_{3}, and C_{4} is
The voltage stress of the output capacitor C_{5} is
The threedimensional surface diagram of the ratio of voltage stress at both ends of the capacitors C_{1}, C_{2}, C_{3}, C_{4} to the output voltage V_{o}, the coupled inductance turns ratio n, and the duty cycle D is shown in Figure 8. It can be seen from Figure 8 that the voltage stress of all capacitors is less than the output voltage if there are different duty cycle D and turns ratio n. The voltage stress of the capacitor C_{4} will reduce significantly if the turns ratio n improves. However, in this case, the voltage stress of the capacitor C_{3} will increase greatly. As duty cycle D increases, the voltage stress of capacitors C_{1} and C_{2} increases gradually, the voltage stress of C_{3} reduces significantly, and the voltage stress of C_{4} increases significantly.
3.2. Comparison of Converter Performance
This section compares the proposed topology with similar highgain nonisolated structures, such as the threewinding coupled inductor converter [6], ASLSU2CCO [8], ASLSU2CVO [9], ZVT converter [10], and the proposed structure in literature [20–24]. The proposed converter is compared with the revealed topologies in the component counts, voltage gain, and the stress of the power switch, as illustrated in Table 1. Under n = 2, the voltage gain curves versus duty cycle and voltage stress on the power switch versus voltage gain are shown in Figures 9 and 10, respectively. The proposed converter can achieve highvoltage gain with lower voltage stress on the power switch.
4. Working Characteristics of Magnetic Components in Converters
4.1. Preselection of Cores
It is known that the output power of the coupled converter is 200 W; the efficiency is assumed to be 80%; the working frequency is 50 kHz; the power loss density per unit volume is 100 mW/cm^{3}; the maximum temperature of the converter is less than 100°C. Choose the core model suitable for the experiment. The design parameters of the converter are presented in Table 2.

The magnetic core in this article is made of material F from the Magnetics company. According to the magnetic core manual, the material has the lowest loss when used at 50°C to 80°C. According to the AP method, the preselected magnetic core formula is as follows:where B_{s} is the saturation magnetic flux density; i_{sp} is the maximum peak current; i_{FL} represents the effective value of the fullload current; K_{1} stands for the product of the current density and window utilization coefficient, and generally the value is 0.0085. There is a strong connection between the effective crosssectional area A_{e} and the maximum power in the magnetic core. is the window area where the magnetic core can wind the wire, which is related to the space of the winding. Therefore, within the allowable space, the larger the product of the two, the better. According to the above formula and the actual application, the model PC36 can be selected for experimental verification. The effective parameters of the magnetic core selected in the experiment are shown in Table 3 [26].

4.2. Design Air Gap Length
According to the working frequency and output power of the experimental design, the potshaped magnetic core is adopted. Considering the convenience of winding manufacturing and the magnetic component installation process, the primary side of the two coupled inductors is tightly wound on the core column, which can better make the secondary side coils closely wound on the surface of the primary side, effectively improving the coupling between the inductors. In order to obtain the required inductance, it is necessary to make the hysteresis loop work in the linear region to avoid saturation of the core and to add an air gap on the core column. The calculation of the air gap is an approximate iterative process. Assuming that the initial air gap value is , a new air gap value can be obtained by substituting it into the air gap formula . The error between the two air gap values is less than the required value, and the iterative calculation ends.
For the selected PC36 magnetic core, D_{cp} is the diameter of the core column; μ_{0} is the permeability in the air. Assuming the initial air gap length , the iterative process is carried out, and the value is obtained by putting the specific data of the magnetic core into formula (17). Compare the values of and , if the error is too large, continue the iteration; when the error is less than 0.4%, stop the iterative process.
4.3. Core Loss Simulation Analysis
In the general calculation of total loss, the loss value is usually ignored, which means that the core loss is roughly regarded as the sum of hysteresis loss and eddy current loss. Figure 11(a) shows the image of the experimental magnetic core, which consists of a cylindrical core of radius r and length l. Here, dr is the differential of the radius r. Assume that the magnetic flux B generated by the current is perpendicular to the crosssection upward. The voltage and current waveforms at both ends of the excitation inductance are shown in Figure 11(b).
(a)
(b)
When sine wave excitation is applied and switches S_{1} and S_{2} are closed, voltage V_{lm} and magnetic induction intensity B_{sin} at both ends of the inductor are follows:
When the duty cycle D = 0.5, under the excitation of the square wave, the relationship between the electromagnetic induction intensity B_{0.5} and the excitation voltage V_{o} can be obtained from the law of electromagnetic induction as follows:
When the duty cycle D changes, so will the magnetic flux density of the core, and the hysteresis loss generated by the magnetic core will also be different. The magnetic flux density amplitude B_{0} is expressed as follows:
Because the hysteresis loss is proportional to the square of the magnetic flux density [13], under the same working conditions, the hysteresis loss caused by different excitation waveforms is the same.
According to formulas (21) and (22), the relationship between the hysteresis loss P_{e} excited by the rectangular wave and the hysteresis loss P_{0.5} excited by the duty cycle D = 0.5 is
According to formulas (22) and (23), the relationship between the hysteresis loss under rectangular wave excitation and the hysteresis loss under sine wave excitation is
From Figure 11(a), it can be seen that the expressions of the instantaneous electromotive force, the instantaneous power and the resistance generated by the magnetic core with radius R are
According to formula (25), the expression of eddy current loss iswhere represents the resistivity of the magnetic materials; A = πr^{2} is the crosssectional area through which the magnetic flux passes; V_{s} is the volume of the magnetic core, which is expressed as πr^{2}l. When the circuit is working in a steady state, using the analysis of the working principle of the converters in the first section and the voltsecond balance, the following expression can be derived.
When the duty cycle of the boost converter and the value of the duty cycle change, the square wave excitation is 0.5 and the expression of the eddy current loss is
Under the same working conditions, the relationship between the rectangular wave excitation eddy current loss P_{e} and the sinusoidal excitation eddy current loss P_{sin} is
When the magnetic core of the converter works in the pulsewidth modulation (PWM) waveform, the bias current I_{dc} will be generated, and the corresponding hysteresis loss will increase significantly. When different excitation waveforms produce the same magnetic flux density under the same conditions, even under the DC bias condition, the hysteresis loss will have a similar value. Therefore, the DC bias factor k_{dc} is proposed to show the effect of DC bias on the hysteresis loss, and k_{dc} is only related to magnetic materials.
According to formulas (24), (29), and (30), the core loss of boost converter under different duty cycle conditions can be expressed as follows:
Ansoft Maxwell is used here for the finite element simulation. Assuming that the parameters of the two coupled inductors are the same and the voltage gain M is the same, the corresponding duty cycle D can be obtained by adjusting the winding turns ratio n of the primary and secondary sides, and the core loss is analyzed under different duty cycles.
4.4. Simulation of Core Loss
The magnetic core parameters are analyzed with a frequency f of 50 kHz, the coupled inductance turns ratio n of 1, and the duty cycle D of 0.5. The magnetic flux density of the magnetic core B_{m} is 0.30884 T and the core loss simulation results are shown in Figure 12.
Secondly, the model is verified and analyzed when the duty cycle is not 0.5. Frequency f is still unchanged, and the value of the turns ratio n in the primary and secondary sides is 2; thus, the core loss with the magnetic flux density is shown in Figure 13. The value of the turns ratio n in the primary and secondary sides is 1.5, the core loss with the magnetic flux density is shown in Figure 14. Table 4 is the analysis and comparison of the theoretical calculation results and the simulation results. It can be seen that the model proposed in this article can accurately evaluate the core loss under different duty cycles.

4.5. Loss Analysis
The analytical loss analysis of the proposed converter is performed by considering the parasitic elements of the components. MOSFETs’losses comprise conduction losses (P_{cond}) and switching losses (P_{sw}), calculated as follows:
The conduction losses of the diodes are calculated by
The conduction losses of the capacitors are calculated by
Coupled inductors losses include copper losses (P_{cu}, CI) and core losses (P_{e}, CI):
The analytical efficiency of the converter is obtained from
When f_{sw} = 50 kHz and existing data are substituted into the formula, the final total loss can be obtained as P_{Loss} = 7.46 W, including P_{S} = 2.96 W, P_{D} = 3.34 W, P_{CI} = 0.36 W, and P_{C} = 0.8 W (the loss of coupling inductance is calculated at duty cycle D = 0.5).
5. Experimental Results and Analysis
5.1. Circuit Experimental Verification
To verify the correctness of the principles in this article, a 200 W experimental prototype was made in the laboratory. The parameters of the experimental prototype are as follows: input voltage V_{in} = 12 V, switching frequency f = 50 kHz, capacitance C_{1} = C_{2} = C_{3} = C_{4} = 100 μF/50 V, C_{5} = 100 μF/250 V, load R = 100 Ω. Coupled inductance turns ratio n is 1. The coupled inductor adopts the tanktype soft ferrite PC36 core. The main circuit parameters are shown in Table 5.

Figure 15 shows the input and output waveforms of the converter. It can be seen that the converter achieves a highgain conversion of 12 V input and 84 V output, which conforms to the theoretical analysis. Figure 16 shows the driving waveforms of the two switch tubes of the converter. Figures 17 and 18, respectively, indicate the voltage stress waveform diagrams of the diode. It can be seen from the figure that the voltage stress of the switch and the diode are less than the output voltage, and the converter has relatively small voltage stress.
Figure 19 shows the current waveforms of the coupled inductor and the switching tube. Figures 20 and 21 show the current waveforms of the main diodes. It can be seen that the diodes have achieved zero current cutoff, and the theoretical analysis is confirmed by all the experimental results.
When the output voltage is constant at 84 V, the efficiency curve of the converter is obtained by changing the power load, as shown in Figure 22. The load can be adjusted and tends to be stable when the efficiency of the prototype keeps increasing. When the output power increases, so does the efficiency of the prototype. Moreover, when the output power is equal to 50 W, the efficiency of the prototype is the highest, about 95.5%.
6. Conclusion
On the basis of the parallel doubleboost structure, a unit with coupled inductance and doubled voltage is proposed, as well as a new type of converters with highgain and lowvoltage stress. Moreover, the voltage gain of the coupled inductance highgain converter in CCM mode, along with the voltage stress of switches and diodes, is derived. Secondly, a core loss model of the converter is formed and then verified through the finite element simulation results. Experimental analyses and verification are done and the converter proposed in this article has the following features:(1)A coupled inductance doubledvoltage unit is introduced into the converter, which greatly improves the voltage gain and simplifies the design difficulty of the converter.(2)The leakage inductance in the coupled inductor affects the parasitic capacitance generated by the converter. The clamping capacitor can absorb the energy of the leakage inductance in the coupled inductor, which can effectively reduce the voltage impulse in the switch tube.(3)Taking ferrite magnetic materials as the research object, while maintaining the same voltage gain, the core loss modeling calculation of the converter is compared with the finite element analysis; the core loss of the converter is evaluated; the model of core loss analysis and modeling is correct.(4)If the converter is used in a special case (PV systems), techniques for reducing commonmode voltage and leakage current must be employed since, in the proposed converter, there is no common ground between the input and output.
Data Availability
The theory of data used to support the findings of this study is included within the article.
Conflicts of Interest
The authors declare that they have no conflicts of interest.
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Copyright © 2021 Hongzhu Li et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.